The present invention relates to a brushless DC motor comprising a rotor having permanent magnets, and more particularly relates to a brushless DC motor capable of reducing cogging torque and a manufacturing method of the same.
A brushless DC motor is a motor that comprises a rotor having permanent magnets and rotates the rotor by controlling an electric commutator circuit for generating a rotational magnetic field in a stator, based on a detection signal representing the rotational position of the rotor. Since the brushless DC motor does not generate mechanical and electrical noises and has high rotary performance and a long life, it is mainly used in the cylinder of a VTR, the capstan of a cassette tape deck, a flexible disk driver, a CD player, etc. In recent years, the brushless DC motor is used in the drive motor of a power steering apparatus for vehicle.
In the brushless DC motor, torque pulsation, i.e., cogging torque, is unavoidably produced because of the presence of slots for winding in the stator and the presence of permanent magnets in the rotor. The cogging torque is a periodical torque change that is caused in a motor by a change of magnetic flux owing to the position of the rotor.
Conventionally, as a method for preventing the cogging torque of a brushless DC motor, there has been a proposed method for reducing cogging torque by dividing the rotor into two blocks and combining the two blocks while displacing the arrangement angle of the rotor blocks in a circumferential direction so that cogging torques generated in the two blocks are mutually in antiphase with respect to the rotation of the rotor.
FIG. 1 is an explanatory view showing an anti-cogging measure taken by such a rotor, and shows a perspective view of a rotor 105 as an assembly of an upper-stage rotor block 110 and a lower-stage rotor block 126. The rotor block 110 comprises an internal rotor core 112, and four permanent magnets 111 attached to the outer circumference of the rotor core 112 at equal intervals. The rotor block 120 comprises an internal rotor core 122, and four permanent magnets 121 attached to the outer circumference of the rotor core 122 at equal intervals.
The rotor blocks 110 and 120 are of the same constructions and combined in an axial direction while displacing the arrangement angles by an amount of a mechanical angle θ12 at which they are mutually in antiphase with respect to a pulsation period of cogging torque generated by the relationship with an opposing stator. Accordingly, the pulsation components of cogging torques generated in the rotor blocks 110 and 120 cancel each other out, thereby reducing the cogging torque of the brushless DC motor.
FIG. 2 is a view showing the relationship between the conventional rotor blocks 110, 120 and the stator as a cross section in a direction perpendicular to the rotary shaft. A stator 101 is formed by layering a number of thin electromagnetic steel plates and fixing them integrally, and comprises a yoke 102 as an outer circumferential portion and teeth 103 that are provided at equal intervals to protrude from the yoke 102 toward the center. Adjacent teeth 103 form a slot 104 together with the yoke 102. Actually, armature windings are wound on the teeth 103 and stored in the slots 104.
FIGS. 3A and 3B are waveform charts for explaining the above-mentioned anti-cogging torque measure. The vertical axis indicates cogging torque, while the horizontal axis shows the rotation angle of the rotor 105. Each of cogging torque Tc1 generated in the upper-stage rotor block 110 and cogging torque Tc2 generated in the lower-stage rotor block 120 has a pulsation period θ11.
In the case where the cogging torques Tc1 and Tc2 are such sinusoidal waveforms that have the same change in an increasing direction and a decreasing direction with respect to the center of amplitude, if the rotor blocks 110 and 120 are combined to have a phase shift corresponding to a half period θ12 of the pulsation period θ11, the pulsation components of the cogging torques Tc1 and Tc2 of the rotor blocks 110 and 120 cancel each other out and, ideally, their composite cogging torque Tct is made a straight waveform having no pulsation as shown in FIG. 3B.
A prerequisite for effectively realizing the above-mentioned method is that the pulsation waveforms of the cogging torques Tc1 and Tc2 generated in the respective rotor blocks 110 and 120 are mutually in antiphase in the moving direction of the rotor with respect to the center of the amplitude and have magnitudes so that the cogging torques Tc1 and Tc2 cancel each other out. In an actual brushless DC motor, it is possible to significantly improve the pulsation period of cogging torque by the above-mentioned method, but there is a problem that small cogging torque pulsation remains. In order to improve such small cogging torque and design a brushless DC motor having no distortion in the rotational operation, conventionally an abrupt change of cogging torque is prevented and the pulsation is restricted by widening the gap between the rotor and the stator to a large extent, providing unequal gaps in the circumferential direction and intentionally leaking a part of magnetic flux from the permanent magnets of the rotor near the region between the magnetic poles. However, the motor efficiency is sacrificed.
FIGS. 4A and 4B are waveform charts for explaining the influence of such an anti-cogging torque measure, in which the same codes as in FIGS. 3A and 3B are used. The pulsation of the cogging torques Tc1 and Tc2 is caused by a change of magnetic flux distribution which occurs between the permanent magnets 111, 121 and the stator 101 with respect to the rotary direction of the rotor 105. In particular, the presence of the openings of the slots 104 of the stator largely affects this change. There is a difference in the magnetic flux distribution in the magnetic path of the stator 101 and the rotor 105 between the case where the region between the magnetic poles of the rotor 105 approaches the opening of the slot 104 and the case where the region between the magnetic poles moves away form the opening. As a result, there is a possibility that the pulsation of cogging torque does not become a sinusoidal waveform having the same change in the increasing direction and the decreasing direction with respect to the center of amplitude.
Moreover, in an ordinary motor structure, the opening of the slot 104 of the stator 101 is narrowed to ensure an interlinkage magnetic flux from the rotor 105 to the stator 101, and the width of the tooth 103 is made larger than the width of the opening of the slot 104 so as to realize high torque and high efficiency. Therefore, the cogging torque has a small change in a section where the region between magnetic poles of the rotor 105 faces the tooth 103 during the rotational movement, but a large change in a section where the region faces the opening of the slot 104. Thus, as shown in FIG. 4A, waveform distortion including even harmonics symmetrical about a point may be caused. Even if the pulsations of such cogging torque waveforms are combined while displacing them by an amount corresponding to the half period 012 of the pulsation period 011, there is a problem that the pulsation of cogging torque as shown in FIG. 4B still remains.
The cogging torque Tct of the rotor obtained by dividing the rotor into two blocks and layering the blocks while displacing them by an amount corresponding to the half period θ12 of the pulsation period θ11 of cogging torque is shown by expression (1). Here, T0 is a peak value of a fundamental wave component of cogging torque when the rotor is not divided, x is an electrical angle of the angle of an arbitrary rotational position of the rotor, n is a natural number, and kn is the ratio of the 2n-th harmonics content to the fundamental wave.                                                         Tct              =                            ⁢                                                TO                  /                  2                                ⁢                                  {                                                            sin                      ⁢                                                                                           ⁢                                              (                        x                        )                                                              +                                                                  ∑                                                  n                          =                          1                                                ∞                                            ⁢                                                                                           ⁢                                                                        kn                          ·                          sin                                                ⁢                                                                                                   ⁢                                                  (                                                      2                            ⁢                            nx                                                    )                                                                                      +                                          sin                      ⁢                                                                                           ⁢                                              (                                                  x                          +                          π                                                )                                                              +                                                                  ∑                                                  n                          =                          1                                                ∞                                            ⁢                                                                                           ⁢                                              kn                        ·                                                                                                                                                                                  ⁢                              sin                ⁢                                                                   ⁢                                  (                                                            2                      ⁢                      nx                                        +                                          2                      ⁢                      n                      ⁢                                                                                           ⁢                      π                                                        )                                            }                                                                          =                            ⁢                              TO                ⁢                                                                   ⁢                                                      ∑                                          n                      =                      1                                        ∞                                    ⁢                                                                           ⁢                                                            kn                      ·                      sin                                        ⁢                                                                                   ⁢                                          (                                              2                        ⁢                        nx                                            )                                                                                                                              (        1        )            
It is apparent from the expression (1) that, in the brushless DC motor comprising the rotor divided into two blocks, the fundamental wave components of cogging torques cancel each other out and are thus eliminated, but there is a problem that the even harmonics components remain.
There is another conventional anti-cogging torque measure shown in FIG. 5, for example. In FIG. 5, the same parts as those shown in FIG. 2 are designated with the same numbers. In the example shown in FIG. 5, one with an outer circumference having a curvature larger than the curvature of the outer circumference of the rotor core 106 is used as the permanent magnet 107 of the rotor 105. Moreover the gap between the permanent magnet 107 and the teeth 103 of the stator 101 gradually increases from the center toward the ends of the permanent magnet 107 in the circumferential direction. Therefore, when the rotor core 106 rotates, the magnetic flux interlinking with the teeth 103 changes smoothly instead of stepwise. Thus, a reduction in cogging torque is made. Some countermeasure produces a similar effect by changing the shape of the outer circumference of the rotor core 106 instead of changing the shape of the permanent magnets 107.
FIG. 6 shows still another conventional anti-cogging torque measure. In the example shown in FIG. 6, a skew angle θS is provided for the arrangement of magnetic poles in the axial direction of the rotor 105. Hence, when the rotor 105 rotates, the timing in which the boundary between the magnetic poles crosses the teeth of the stator varies according to a position in the axial direction of the rotor 105. Thus, the change of the magnetic flux interlinking with the teeth is made moderate, and the cogging torque is reduced.
However, both of the conventional techniques shown in FIGS. 5 and 6 suffer from problems including poor magnetic efficiency. In the technique shown in FIG. 5, since the average gap between the permanent magnets 107 and the teeth 103 is large, the magnetic efficiency is poor and a rotary output proportional to the magnetic force of the permanent magnets 107 can not be obtained. Moreover, it is necessary to perform various analysis and trial manufacture to determine the shape of the permanent magnets 107 or the outer circumference of the rotor core 106, resulting in high development costs. Furthermore, it is necessary to process the small configurations accurately, and thus the processing itself is difficult. Nevertheless, an objective to reduce the cogging torque is not sufficiently achieved. In particular, when strong rare-earth based permanent magnets are used to meet the demand for a reduction in size as in recent years, the cogging torque in itself is considerably large. Therefore, such a method is not sufficient.
Similarly, the technique shown in FIG. 6 suffers from poor magnetic efficiency and can not obtain a sufficient rotary output. The reason for this is that there is the skew angle θS in the arrangement of magnetic poles and consequently the effective magnetic flux of the magnetic poles becomes smaller by a corresponding amount. In the example shown in FIG. 6, one magnetic pole occupies substantially a parallelogram region on the side face of the rotor 105. In a portion near the acute apex, the magnetic flux in the portion does not effectively perform the function of the motor. Therefore, like the example shown in FIG. 5, this technique can not obtain a sufficient rotary output.
In recent years, the brushless DC motor is often made to have a small size and high output by using a rare-earth material, etc. for the permanent magnets, and tends to be used as a magnetic circuit in a high magnetic flux density region of the thin electromagnetic steel plates. On the other hand, there is a problem that the motor performance is degraded as a result of the promotion of the reduction in the size of the motor and the generation of extremely high heat by the motor for the size of the motor. In order to solve this problem, notch portions are provided in the outer circumference of the stator and a cool air or the like is caused to flow through the notch portions to cool the motor and limit the generation of heat. Besides, in order to achieve another objective to ensure a punching yield of electromagnetic steel plates and a gap in the layering direction for sticking means such as welding, notch portions are provided on the outer circumference side of the stator.
FIG. 7 is a perspective view showing an example of the stator of a conventional brushless DC motor having such notch portions. In FIG. 7, the same parts as in those of FIGS. 2 and 5 are designated with the same numbers.
A notch portion 109 running from the upper end to the lower end of the stator 101 is provided on the outer circumferential surface of the yoke 102, at a position near the outside of every third tooth 103. The notch portions 109 are provided on the outer circumference as the cooling paths for releasing heat during the operation of the motor and for the purpose of easing the welding that is performed for fixing plural layered steel plates (by using the protrusions in the notch portions 109) and easing the punching of material to improve the yield. By providing the notch portions 109 on the outer circumference, it is possible to prevent the welded section from fixing out of the outer circumference of the stator 101 in welding the thin electromagnetic thin plates to fix them integrally. Moreover, the notch portions 109 are often provided for the purpose of saving the material of the thin electromagnetic steel plates of the stator 101. As described above, each of the notch portions 109 runs from the upper end to the lower end of the stator 101 and has a length S0 in the layering direction.
In the above-described conventional stator 101, since the notch portions 109 are aligned with the layering direction, there is a difference in the magnetic resistance seen from the inside of the stator 101 between a region of the teeth 103 where the notch portion 109 is present on the outer circumference side of the stator 101 and a region where the notch portion 109 is not present. In the case where a rotor having permanent magnets is positioned inside the stator 101, a magnetic circuit in which the magnetic flux flows is formed between the stator 101 and the rotor which faces the teeth 103 and have permanent magnets arranged so that adjacent permanent magnetic have opposite polarities. This magnetic circuit is formed as a magnetic closed circuit composed mainly of the shortest path between adjacent opposite poles. The shortest magnetic circuit starting from a region between the magnetic poles of the permanent magnets of the rotor is most of the causes of generation of cogging torque. In this magnetic circuit, there is a big difference in the magnetic flux amount between a region of the teeth 103 where the notch portion 109 is present on the outer circumference side of the stator 101 and a region where the notch portion 109 is not present. Thus, the difference in the magnetic flux amount according to the positions in the rotary direction of the rotor is one of the causes of cogging torque, and is a cause of generation of sound and vibration.
FIG. 8 is a view showing the state of magnetic flux in such a brushless DC motor. Here, the rotor 105 having the permanent magnets 107 attached to the surface of the rotor core 106 is disposed inside the stator 101 shown in FIG. 7. The stator 101 is formed by layering necessary pieces of thin electromagnetic steel plates having a portion equivalent to the notch portion 109 on the outer circumference side of every third portion equivalent to the tooth 103. Note that the rotor 105 may be a buried-type rotor having permanent magnets buried in the rotor core 106.
A magnetic flux generated by the relative positional relationship between the stator 101 and the regions between the magnetic poles of the opposing permanent magnets 107 of the rotor 105 flows in respective portions of the stator 101. The magnetic flux amount in a magnetic path a in a region of the teeth 103 where a notch portion 109 is present on the outer circumference side of the stator 101 is denoted as φ1, the magnetic flux amount in a magnetic path b in a region of the teeth 103 where no notch portion 109 is present on the outer circumference side of the stator 101 is denoted as φ2, and the magnetic flux amount in a magnetic path c in a region where a notch portion 109 different from that for the flux amount φ1 is present is denoted as φ3. Here, if the notch portions 109 have the same configuration, it is clear that only the difference between the magnetic flux amounts φ1 and φ3 is the position of the notch portion 109 in the magnetic path, and the magnetic flux amounts φ1 and φ3 are the same in magnitude.
Here, as shown in FIG. 8, when straight lines A, B and C are drawn from the center of the shaft hole of the rotor 105 through the center of the slots 104 toward the outer circumference of the stator 101, if a region between the magnetic poles of the permanent magnets 107 of the rotor 105 is positioned on the straight line A, the magnetic flux from the permanent magnets 107 near the region between the magnetic poles forms a closed circuit of the magnetic flux amount φ1 by the magnetic path a shown by a dotted line. Besides, when the rotor 105 rotates clockwise and the region between the magnetic poles of the permanent magnets 107 reaches the straight line B, the magnetic flux from the permanent magnets 107 near the region between the magnetic poles forms a closed circuit of the flux amount φ2 by the magnetic path b shown by a dotted line. When the rotor 105 further rotates clockwise and the region between the magnetic poles of the permanent magnets 107 reaches the straight line C, the magnetic flux from the permanent magnets 107 near the region between the magnetic poles forms a closed circuit of the flux amount φ3 by the magnetic path c shown by a dotted line.
There is a difference in the cross sectional area of the magnetic path due to the presence and absence of the notch portion 109 in the magnetic path, between the state where the region between the magnetic poles of the permanent magnets 107 of the rotor 105 is positioned on the straight line A and the state where the region between the magnetic poles is positioned on the straight line B. Accordingly, there is a difference in the magnetic resistance, and the flux amounts are φ1<φ2. Similarly, there is a difference in the cross sectional area of the magnetic path due to the presence and absence of the notch portion 109, between the state where the region between the magnetic poles of the permanent magnets 107 of the rotor 105 is positioned on the straight line B and the state where the region between the magnetic poles is positioned on the straight line C. Accordingly, there is a difference in the magnetic resistance, and the flux amounts are φ3<φ2.
Hence, when the region between the magnetic poles of the permanent magnets 107 of the rotor 105 is positioned on the straight line B having no notch portion 109 on the outer circumference side of the stator 101, the strongest magnetic coupling is obtained between the rotor 105 and the stator 101. The change in cogging torque resulting from such phenomena is that the largest cogging torque appears when the region between the magnetic poles approaches or moves away from the position of the straight line B because the magnetic coupling is strong in that position as shown in FIG. 9 and described above. In FIG. 9, the vertical axis indicates the cogging torque TC and the horizontal axis shows the rotation angle θ of the rotor 105, and the positions of the straight lines A to C shown in FIG. 8 correspond to the positions of the straight lines A to C of FIG. 9.
However, in the brushless DC motor, since the notch portions 109 are provided on the outer circumference side of the stator 101, the size of the cross sectional area of the magnetic paths varies because of the difference in the magnetic paths as described above. As a result, the brushless DC motor suffers from a problem of deterioration of the pulsation of cogging torque.